1
LTC1872
APPLICATIO S
U
FEATURES
DESCRIPTIO
U
TYPICAL APPLICATIO
N
U
Constant Frequency
Current Mode Step-Up
DC/DC Controller in SOT-23
The LTC
1872 is a constant frequency current mode step-
up DC/DC controller providing excellent AC and DC load
and line regulation. The device incorporates an accurate
undervoltage lockout feature that shuts down the LTC1872
when the input voltage falls below 2.0V.
The LTC1872 boasts a
2.5% output voltage accuracy and
consumes only 270
A of quiescent current. For applica-
tions where efficiency is a prime consideration, the LTC1872
is configured for Burst Mode operation, which enhances
efficiency at low output current.
In shutdown, the device draws a mere 8
A. The high
550kHz constant operating frequency allows the use of a
small external inductor.
The LTC1872 is available in a small footprint 6-lead
SOT-23.
s
Lithium-Ion-Powered Applications
s
Cellular Telephones
s
Wireless Modems
s
Portable Computers
s
Scanners
s
High Efficiency: Over 90%
s
High Output Currents Easily Achieved
s
Wide V
IN
Range: 2.5V to 9.8V
s
V
OUT
Limited Only by External Components
s
Constant Frequency 550kHz Operation
s
Burst Mode
TM
Operation at Light Load
s
Current Mode Operation for Excellent Line and Load
Transient Response
s
Low Quiescent Current: 270
A
s
Shutdown Mode Draws Only 8
A Supply Current
s
2.5% Reference Accuracy
s
Tiny 6-Lead SOT-23 Package
Efficiency vs Load Current
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
I
TH
/RUN
LTC1872
147k
422k
80.6k
R1
0.03
L1
4.7
H
220pF
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: MURATA GRM42-2X5R226K6.3
D1: IR10BQ015
L1: MURATA LQN6C4R7M04
M1: IRLMS2002
R1: DALE 0.25W
GND
V
FB
5
4
6
1872 TA01
1
2
3
NGATE
V
IN
SENSE
C1
10
F
10V
V
IN
3.3V
V
OUT
5V
1A
C2
2
22
F
6.3V
+
M1
D1
Figure 1. LTC1872 High Output Current 3.3V to 5V Boost Converter
LOAD CURRENT (mA)
1
EFFICIENCY (%)
100
95
90
85
80
75
70
65
10
100
1000
1872 TA01b
V
IN
= 3.3V
V
OUT
= 5V
2
LTC1872
ABSOLUTE
M
AXI
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RATINGS
W
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(Note 1)
Input Supply Voltage (V
IN
)......................... 0.3V to 10V
SENSE
, NGATE Voltages ............ 0.3V to (V
IN
+ 0.3V)
V
FB
, I
TH
/RUN Voltages .............................. 0.3V to 2.4V
NGATE Peak Output Current (< 10
s) ....................... 1A
Storage Ambient Temperature Range ... 65
C to 150
C
Operating Temperature Range (Note 2) .. 40
C to 85
C
Junction Temperature (Note 3) ............................. 150
C
Lead Temperature (Soldering, 10 sec).................. 300
C
PACKAGE/ORDER I
N
FOR
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ATIO
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T
JMAX
= 150
C,
JA
= 230
C/ W
S6 PART MARKING
ORDER PART
NUMBER
LTC1872ES6
LTMK
Consult factory for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The
q
denotes specifications that apply over the full operating temperature
range, otherwise specifications are at T
A
= 25
C. V
IN
= 4.2V unless otherwise specified. (Note 2)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input DC Supply Current
Typicals at V
IN
= 4.2V (Note 4)
Normal Operation
2.4V
V
IN
9.8V
270
420
A
Sleep Mode
2.4V
V
IN
9.8V
230
370
A
Shutdown
2.4V
V
IN
9.8V, V
ITH
/RUN = 0V
8
22
A
UVLO
V
IN
< UVLO Threshold
6
10
A
Undervoltage Lockout Threshold
V
IN
Falling
q
1.55
2.00
2.35
V
V
IN
Rising
1.85
2.10
2.40
V
Shutdown Threshold (at I
TH
/RUN)
q
0.15
0.35
0.55
V
Start-Up Current Source
V
ITH
/RUN = 0V
0.25
0.5
0.85
A
Regulated Feedback Voltage
0
C to 70
C(Note 5)
q
0.780
0.800
0.820
V
40
C to 85
C(Note 5)
q
0.770
0.800
0.830
V
V
FB
Input Current
(Note 5)
10
50
nA
Oscillator Frequency
V
FB
= 0.8V
500
550
650
kHz
Gate Drive Rise Time
C
LOAD
= 3000pF
40
ns
Gate Drive Fall Time
C
LOAD
= 3000pF
40
ns
Peak Current Sense Voltage
(Note 6)
114
120
mV
I
TH
/RUN 1
GND 2
V
FB
3
6 NGATE
5 V
IN
4 SENSE
TOP VIEW
S6 PACKAGE
6-LEAD PLASTIC SOT-23
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC1872E is guaranteed to meet performance specifications
from 0
C to 70
C. Specifications over the 40
C to 85
C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: T
J
is calculated from the ambient temperature T
A
and power
dissipation P
D
according to the following formula:
T
J
= T
A
+ (P
D
JA
C/W)
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 5: The LTC1872 is tested in a feedback loop that servos V
FB
to the
output of the error amplifier.
Note 6: Guaranteed by design at duty cycle = 30%. Peak current sense
voltage is V
REF
/6.67 at duty cycle <40%, and decreases as duty cycle
increases due to slope compensation as shown in Figure 2.
3
LTC1872
TYPICAL PERFOR
M
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CE CHARACTERISTICS
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Reference Voltage
vs Temperature
Undervoltage Lockout Trip
Voltage vs Temperature
Shutdown Threshold
vs Temperature
Maximum Current Sense Trip
Voltage vs Duty Cycle
Normalized Oscillator Frequency
vs Temperature
PI
N
FU
N
CTIO
N
S
U
U
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I
TH
/RUN (Pin 1): This pin performs two functions. It
serves as the error amplifier compensation point as well as
the run control input. Nominal voltage range for this pin is
0.7V to 1.9V. Forcing this pin below 0.35V causes the
device to be shut down. In shutdown all functions are
disabled and the NGATE pin is held low.
GND (Pin 2): Ground Pin.
V
FB
(Pin 3): Receives the feedback voltage from an exter-
nal resistive divider across the output.
SENSE
(Pin 4): The Negative Input to the Current Com-
parator.
V
IN
(Pin 5): Supply Pin. Must be closely decoupled to GND
Pin 2.
NGATE (Pin 6): Gate Drive for the External N-Channel
MOSFET. This pin swings from 0V to V
IN
.
TEMPERATURE (
C)
55
775
V
FB
VOLTAGE (mV)
780
790
795
800
825
810
15
25
45
125
1872 G01
785
815
820
805
35
5
65
85 105
V
IN
= 4.2V
TEMPERATURE (
C)
55
10
NORMALIZED FREQUENCY (%)
8
4
2
0
10
4
15
25
45
125
1872 G02
6
6
8
2
35
5
65
85 105
V
IN
= 4.2V
TEMPERATURE (
C)
55
1.84
UVLO TRIP VOLTAGE (V)
1.88
1.96
2.00
2.04
2.24
2.12
15
25
45
125
1872 G03
1.92
2.16
2.20
2.08
35
5
65
85 105
V
IN
FALLING
DUTY CYCLE (%)
20
30
V
IN
V
SENSE
(mV)
100
1872 G04
40
50
60
70
80
90
130
120
110
100
90
80
70
60
50
V
IN
= 4.2V
T
A
= 25
C
TEMPERATURE (
C)
55
200
I
TH
/RUN VOLTAGE (mV)
240
320
360
400
600
480
15
25
45
125
1872 G05
280
520
560
440
35
5
65
85 105
V
IN
= 4.2V
4
LTC1872
FU
N
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AL DIAGRA
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OPERATIO
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(Refer to Functional Diagram)
Main Control Loop
The LTC1872 is a constant frequency current mode switch-
ing regulator. During normal operation, the external
N-channel power MOSFET is turned on each cycle by the
oscillator and turned off when the current comparator
(ICMP) resets the RS latch. The peak inductor current at
which ICMP resets the RS latch is controlled by the voltage
on the I
TH
/RUN pin, which is the output of the error
amplifier EAMP. An external resistive divider connected
between V
OUT
and ground allows the EAMP to receive an
output feedback voltage V
FB
. When the load current in-
creases, it causes a slight decrease in V
FB
relative to the
0.8V reference, which in turn causes the
I
TH
/RUN voltage to increase until the average inductor
current matches the new load current.
The main control loop is shut down by pulling the I
TH
/RUN
pin low. Releasing I
TH
/RUN allows an internal 0.5
A
current source to charge up the external compensation
network. When the I
TH
/RUN pin reaches 0.35V, the main
control loop is enabled with the I
TH
/RUN voltage then
pulled up to its zero current level of approximately 0.7V.
As the external compensation network continues to charge
up, the corresponding output current trip level follows,
allowing normal operation.
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
+
+
+
0.15V
0.5
A
0.3V
SLEEP
OVP
BURST
CMP
SHDN
1.2V
UV
1872FD
V
REF
+
60mV
V
REF
0.8V
V
IN
RS
VOLTAGE
REFERENCE
SLOPE
COMP
ICMP
R
S
Q
FREQ
FOLDBACK
OSC
SENSE
V
IN
5
2
4
+
4
EAMP
V
FB
+
3
NGATE
V
IN
6
I
TH
/RUN
V
IN
0.35V
V
REF
0.8V
1
+
SHDN
CMP
0.3V
GND
+
UNDERVOLTAGE
LOCKOUT
V
IN
5
LTC1872
OPERATIO
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(Refer to Functional Diagram)
Comparator OVP guards against transient overshoots
> 7.5% by turning off the external N-channel power
MOSFET and keeping it off until the fault is removed.
Burst Mode Operation
The LTC1872 enters Burst Mode operation at low load
currents. In this mode, the peak current of the inductor is
set as if V
ITH
/RUN = 1V (at low duty cycles) even though
the voltage at the I
TH
/RUN pin is at a lower value. If the
inductor's average current is greater than the load require-
ment, the voltage at the I
TH
/RUN pin will drop. When the
I
TH
/RUN voltage goes below 0.85V, the sleep signal goes
high, turning off the external MOSFET. The sleep signal
goes low when the I
TH
/RUN voltage goes above 0.925V
and the LTC1872 resumes normal operation. The next
oscillator cycle will turn the external MOSFET on and the
switching cycle repeats.
Undervoltage Lockout
To prevent operation of the N-channel MOSFET below safe
input voltage levels, an undervoltage lockout is incorpo-
rated into the LTC1872. When the input supply voltage
drops below approximately 2.0V, the N-channel MOSFET
and all circuitry is turned off except the undervoltage
block, which draws only several microamperes.
Figure 2. Maximum Output Current vs Duty Cycle
DUTY CYCLE (%)
110
100
90
80
70
60
50
40
30
20
10
SF = I
OUT
/I
OUT(MAX)
(%)
1872 F02
0
70 80 90 100
60
10
20 30
40 50
I
RIPPLE
= 0.4I
PK
AT 5% DUTY CYCLE
I
RIPPLE
= 0.2I
PK
AT 5% DUTY CYCLE
V
IN
= 4.2V
Overvoltage Protection
The overvoltage comparator in the LTC1872 will turn the
external MOSFET off when the feedback voltage has risen
7.5% above the reference voltage of 0.8V. This compara-
tor has a typical hysteresis of 20mV.
Slope Compensation and Inductor's Peak Current
The inductor's peak current is determined by:
I
V
R
PK
ITH
SENSE
=
-
(
)
0 7
10
.
when the LTC1872 is operating below 40% duty cycle.
However, once the duty cycle exceeds 40%, slope com-
pensation begins and effectively reduces the peak induc-
tor current. The amount of reduction is given by the curves
in Figure 2.
Short-Circuit Protection
Since the power switch in a boost converter is not in series
with the power path from input to load, turning off the
switch provides no protection from a short-circuit at the
output. External means such as a fuse in series with the
boost inductor must be employed to handle this fault
condition.
6
LTC1872
APPLICATIO
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The basic LTC1872 application circuit is shown in
Figure 1. External component selection is driven by the
load requirement and begins with the selection of L1 and
R
SENSE
(= R1). Next, the power MOSFET and the output
diode D1 is selected followed by C
IN
(= C1) and C
OUT
(= C2).
R
SENSE
Selection for Output Current
R
SENSE
is chosen based on the required output current.
With the current comparator monitoring the voltage devel-
oped across R
SENSE
, the threshold of the comparator
determines the inductor's peak current. The output cur-
rent the LTC1872 can provide is given by:
I
R
I
V
V
V
OUT
SENSE
RIPPLE
IN
OUT
D
=
-
+
0 12
2
.
where I
RIPPLE
is the inductor peak-to-peak ripple current
(see Inductor Value Calculation section) and V
D
is the
forward drop of the output diode at the full rated output
current.
A reasonable starting point for setting ripple current is:
I
O
I
V
V
V
RIPPLE
OUT
OUT
D
IN
=
( )( )
+
.4
Rearranging the above equation, it becomes:
R
I
V
V
SENSE
OUT
OUT
D
=
( )( )
+
1
10
V
for Duty Cycle < 40%
IN
However, for operation that is above 40% duty cycle, slope
compensation's effect has to be taken into consideration
to select the appropriate value to provide the required
amount of current. Using the scaling factor (SF, in %) in
Figure 2, the value of R
SENSE
is:
R
SF
I
V
V
V
SENSE
OUT
IN
OUT
D
=
( )( )( )
+
10
100
Inductor Value Calculation
The operating frequency and inductor selection are inter-
related in that higher operating frequencies permit the use
of a smaller inductor for the same amount of inductor
ripple current. However, this is at the expense of efficiency
due to an increase in MOSFET gate charge losses.
The inductance value also has a direct effect on ripple
current. The ripple current, I
RIPPLE
, decreases with higher
inductance or frequency and increases with higher V
OUT
.
The inductor's peak-to-peak ripple current is given by:
I
V
f L
V
V
V
V
V
RIPPLE
IN
OUT
D
IN
OUT
D
=
( )
+
-
+
where f is the operating frequency. Accepting larger values
of I
RIPPLE
allows the use of low inductances, but results in
higher output voltage ripple and greater core losses. A
reasonable starting point for setting ripple current is:
I
I
V
V
V
RIPPLE
OUT MAX
OUT
D
IN
=
+
( )
0 4
.
In Burst Mode operation, the ripple current is normally set
such that the inductor current is continuous during the
burst periods. Therefore, the peak-to-peak ripple current
must not exceed:
I
R
RIPPLE
SENSE
0 03
.
This implies a minimum inductance of:
L
V
f
R
V
V
V
V
V
MIN
IN
SENSE
OUT
D
IN
OUT
D
=
+
-
+
0 03
.
A smaller value than L
MIN
could be used in the circuit;
however, the inductor current will not be continuous
during burst periods.
7
LTC1872
APPLICATIO
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Inductor Selection
When selecting the inductor, keep in mind that inductor
saturation current has to be greater than the current limit
set by the current sense resistor. Also, keep in mind that
the DC resistance of the inductor will affect the efficiency.
Off the shelf inductors are available from Murata, Coilcraft,
Toko, Panasonic, Coiltronics and many other suppliers.
Power MOSFET Selection
The main selection criteria for the power MOSFET are the
threshold voltage V
GS(TH)
, the "on" resistance R
DS(ON)
,
reverse transfer capacitance C
RSS
and total gate charge.
Since the LTC1872 is designed for operation down to low
input voltages, a logic level threshold MOSFET (R
DS(ON)
guaranteed at V
GS
= 2.5V) is required for applications that
work close to this voltage. When these MOSFETs are used,
make sure that the input supply to the LTC1872 is less than
the absolute maximum V
GS
rating, typically 8V.
The required minimum R
DS(ON)
of the MOSFET is gov-
erned by its allowable power dissipation given by:
R
P
DC I
p
DS ON
P
IN
(
)
( )
+
( )
2
1
where P
P
is the allowable power dissipation and
p is the
temperature dependency of R
DS(ON)
. (1 +
p) is generally
given for a MOSFET in the form of a normalized R
DS(ON)
vs
temperature curve, but
p = 0.005/
C can be used as an
approximation for low voltage MOSFETs. DC is the maxi-
mum operating duty cycle of the LTC1872.
Output Diode Selection
Under normal load conditions, the average current con-
ducted by the diode in a boost converter is equal to the
output load current:
I
I
D avg
OUT
(
)
=
It is important to adequately specify the diode peak current
and average power dissipation so as not to exceed the
diode ratings.
Schottky diodes are recommended for low forward drop
and fast switching times. Remember to keep lead length
short and observe proper grounding (see Board Layout
Checklist) to avoid ringing and increased dissipation.
C
IN
and C
OUT
Selection
To prevent large input voltage ripple, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current for a boost
converter is approximately equal to:
C
I
IN
RIPPLE
Required I
RMS
( )
0 3
.
where I
RIPPLE
is as defined in the Inductor Value Calcula-
tion section.
Note that capacitor manufacturer's ripple current ratings
are often based on 2000 hours of life. This makes it
advisable to further derate the capacitor, or to choose a
capacitor rated at a higher temperature than required.
Several capacitors may be paralleled to meet the size or
height requirements in the design. Due to the high operat-
ing frequency of the LTC1872, ceramic capacitors can also
be used for C
IN
. Always consult the manufacturer if there
is any question.
The selection of C
OUT
is driven by the required effective
series resistance (ESR). Typically, once the ESR require-
ment is satisfied, the capacitance is adequate for filtering.
The output ripple (
V
OUT
) is approximated by:
V
I
V
V
V
I
ESR
f C
OUT
O
OUT
D
IN
RIPPLE
OUT
+
+
+
2
1
2
2
2
1
2
8
LTC1872
Figure 4. Setting Output Voltage
3
V
FB
V
OUT
LTC1872
R1
1872 F04
R2
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Setting Output Voltage
The LTC1872 develops a 0.8V reference voltage between
the feedback (Pin 3) terminal and ground (see Figure 4). By
selecting resistor R1, a constant current is caused to flow
through R1 and R2 to set the overall output voltage. The
regulated output voltage is determined by:
V
V
R
R
OUT
=
+
0 8
1
2
1
.
where f is the operating frequency, C
OUT
is the output
capacitance and I
RIPPLE
is the ripple current in the induc-
tor.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance through-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat
higher price. The output capacitor RMS current is approxi-
mately equal to:
I
DC
DC
PK
-
2
where I
PK
is the peak inductor current and DC is the switch
duty cycle.
When using electrolytic output capacitors, if the ripple and
ESR requirements are met, there is likely to be far more
capacitance than required.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
surface mount configurations. An excellent choice of
tantalum capacitors is the AVX TPS and KEMET T510
series of surface mount tantalum capacitors. Also,
ceramic capacitors in X5R pr X7R dielectrics offer excel-
lent performance.
Low Supply Operation
Although the LTC1872 can function down to approxi-
mately 2.0V, the maximum allowable output current is
reduced when V
IN
decreases below 3V. Figure 3 shows the
amount of change as the supply is reduced down to 2V.
Also shown in Figure 3 is the effect of V
IN
on V
REF
as V
IN
goes below 2.3V.
Figure 3. Line Regulation of V
REF
and V
ITH
INPUT VOLTAGE (V)
2.0
NORMALIZED VOLTAGE (%)
105
100
95
90
85
80
75
2.2
2.4
2.6
2.8
1872 F03
3.0
V
REF
V
ITH
9
LTC1872
For most applications, an 80k resistor is suggested for R1.
To prevent stray pickup, locate resistors R1 and R2 close
to LTC1872.
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% (
1 +
2 +
3 + ...)
where
1,
2, etc. are the individual losses as a percent-
age of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1872 circuits: 1) LTC1872 DC bias current,
2) MOSFET gate charge current, 3) I
2
R losses and 4)
voltage drop of the output diode.
1. The V
IN
current is the DC supply current, given in the
electrical characteristics, that excludes MOSFET driver
and control currents. V
IN
current results in a small loss
which increases with V
IN
.
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2. MOSFET gate charge current results from switching
the gate capacitance of the power MOSFET. Each time
a MOSFET gate is switched from low to high to low
again, a packet of charge, dQ, moves from V
IN
to
ground. The resulting dQ/dt is a current out of V
IN
which is typically much larger than the contoller's DC
supply current. In continuous mode, I
GATECHG
= f(Qp).
3. I
2
R losses are predicted from the DC resistances of the
MOSFET, inductor and current sense resistor. The
MOSFET R
DS(ON)
multiplied by duty cycle times the
average output current squared can be summed with
I
2
R losses in the inductor ESR in series with the current
sense resistor.
4. The output diode is a major source of power loss at
high currents. The diode loss is calculated by multiply-
ing the forward voltage by the load current.
5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
Transition Loss = 2(V
IN
)
2
I
IN(MAX)
C
RSS
(f)
Other losses, including C
IN
and C
OUT
ESR dissipative
losses, and inductor core losses, generally account for
less than 2% total additional loss.
10
LTC1872
APPLICATIO
N
S I
N
FOR
M
ATIO
N
W
U
U
U
Figure 5. LTC1872 Layout Diagram (See PC Board Layout Checklist)
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1872. These items are illustrated graphically in the
layout diagram in Figure 5. Check the following in your
layout:
1. The Schottky diode should be closely connected
between the output capacitor and the drain of the
external MOSFET.
2. The (+) plate of C
IN
should connect to the sense
resistor as closely as possible. This capacitor provides
AC current to the inductor.
3. The input decoupling capacitor (0.1
F) should be
connected closely between V
IN
(Pin 5) and ground
(Pin 2).
4. Connect the end of R
SENSE
as close to V
IN
(Pin 5) as
possible. The V
IN
pin is the SENSE
+
of the current
comparator.
5. The trace from SENSE
(Pin 4) to the Sense resistor
should be kept short. The trace should connect close
to R
SENSE
.
6. Keep the switching node NGATE away from sensitive
small signal nodes.
7. The V
FB
pin should connect directly to the feedback
resistors. The resistive divider R1 and R2 must be
connected between the (+) plate of C
OUT
and signal
ground.
L1
M1
BOLD LINES INDICATE HIGH CURRENT PATHS
D1
R
S
V
OUT
V
IN
1872 F05
0.1
F
C
ITH
R
ITH
R2
R1
C
IN
+
I
TH
/RUN
LTC1872
GND
V
FB
6
5
4
1
2
3
NGATE
V
IN
SENSE
C
OUT
+
11
LTC1872
LTC1872 12V/500mA Boost Converter
TYPICAL APPLICATIO
U
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
LTC1872 Three-Cell White LED Driver
I
TH
/RUN
LTC1872
10k
R1
0.033
1.1M
78.7k
L1
10
H
220pF
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: AVX TPSE476M016R0150
D1: IR10BQ015
L1: COILTRONICS UP2B-100
M1: Si9804DV
R1: DALE 0.25W
GND
V
FB
5
4
6
1872 TA02
1
2
3
NGATE
V
IN
SENSE
C1
10
F
10V
V
IN
3V TO 9.8V
V
OUT
12V
C2
47
F
16V
+
M1
D1
I
TH
/RUN
LTC1872
10k
R1
0.27
53.6
L1
150
H
220pF
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: AVX TPSD156M035R0300
D0: MOTOROLA MBR0540
D1-D7: CMD333UWC
GND
V
FB
5
4
6
1872 TA04
1
2
3
NGATE
V
IN
SENSE
C1
10
F
10V
V
OUT
28.8V
(WITH 8 LEDs)
C2
15
F
35V
+
C3
0.1
F
CERAMIC
M1
D0
1 TO 8
WHITE
LEDs
15mA
D1
D2
D8
AA
AA
AA
V
IN
= 3 AA CELLS
2.7V TO 4.8V
L1: COILCRAFT DO1608C-154
M1: Si9804
R1: DALE 0.25W
12
LTC1872
1872f LT/TP 0301 4K PRINTED IN USA
PART NUMBER
DESCRIPTION
COMMENTS
LT1304
Micropower DC/DC Converter with Low-Battery Detector
120
A Quiescent Current, 1.5V
V
IN
8V
LT1610
1.7MHz, Single Cell Micropower DC/DC Converter
30
A Quiescent Current, V
IN
Down to 1V
LT1613
1.4MHz, Single Cell DC/DC Converter in 5-Lead SOT-23
Internally Compensated, V
IN
Down to 1V
LT1619
Low Voltage Current Mode PWM Controller
8-Lead MSOP Package, 1.9V
V
IN
18V
LT1680
High Power DC/DC Step-Up Controller
Operation Up to 60V, Fixed Frequency Current Mode
LTC1624
High Efficiency SO-8 N-Channel Switching Regulator Controller
8-Pin N-Channel Drive, 3.5V
V
IN
36V
LT1615
Micropower Step-Up DC/DC Converter in SOT-23
20
A Quiescent Current, V
IN
Down to 1V
LTC1700
No R
SENSE
Synchronous Current Mode DC/DC Step-Up Controller
95% Efficient, 0.9V
V
IN
5V, 550kHz Operation
LTC1772
Constant Frequency Current Mode Step-Down DC/DC Controller
V
IN
2.5V to 9.8V, I
OUT
up to 4A, SOT-23 Package
LTC3401/LTC3402
1A/2A, 3MHz Micropower Synchronous Boost Converter
10-Lead MSOP Package, 0.5V
V
IN
5V
LINEAR TECHNOLOGY CORPORATION 2000
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
q
FAX: (408) 434-0507
q
www.linear-tech.com
TYPICAL APPLICATIO
U
LTC1872 2.5V to 3.3V/0.5A Boost Converter
RELATED PARTS
I
TH
/RUN
LTC1872
R
C1
10k
R
CS
0.03
L1A
L1B
C
C1
220pF
GND
V
FB
5
4
6
1872 TA05
1
2
3
NGATE
V
IN
SENSE
C
IN
10
F
10V, X5R
V
IN
2.7V TO 9.8V
V
OUT
3.3V/1.2A
C01
180
F
4V, SP
+
M1
CS
4.7
F
10V
D1
MBRM120
R
f1
252k
R
f2
80.6k
FOR V
OUT
= 5V CHANGE
R
f1
TO 427k
AND
C
01
TO 150
F, 6V PANASONIC
SP TYPE CAPACITOR
C
IN
, CS; TOKO, MURATA OR TAIYO YUDEN
C
01
: PANASONIC EEFUE0G181R
L1: BH ELECTRONICS 511-1012
M1: IRLMS2002
R
CS
: DALE OR IRC
I
TH
/RUN
LTC1872
10k
R1
0.034
332k
U1
80.6k
180k
L1
4.7
H
220pF
0.1
F
CERAMIC
C1, C2: AVX TPSE107M010R0100
D1: MOTOROLA MBR2045CT
L1: COILTRONICS UP2B-4R7
M1: Si9804DV
R1: DALE 0.25W
U1: PANASONIC 2SB709A
GND
V
FB
5
4
6
1872 TA03
1
2
3
NGATE
V
IN
SENSE
C2
2
100
F
10V
V
IN
2.5V
V
OUT
3.3V
0.5A
M1 D1
+
C1
100
F
10V
+
LTC1872 2.7V to 9.8V Input
to 3.3V/1.2A Output SEPIC Converter
Dimensions in inches (millimeters) unless otherwise noted.
PACKAGE DESCRIPTIO
N
U
0.95
(0.037)
REF
1.50 1.75
(0.059 0.069)
0.35 0.55
(0.014 0.022)
0.35 0.50
(0.014 0.020)
SIX PLACES (NOTE 2)
S6 SOT-23 0898
2.80 3.00
(0.110 0.118)
(NOTE 3)
1.90
(0.074)
REF
0.90 1.45
(0.035 0.057)
0.90 1.30
(0.035 0.051)
0.00 0.15
(0.00 0.006)
0.09 0.20
(0.004 0.008)
(NOTE 2)
2.6 3.0
(0.110 0.118)
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DIMENSIONS ARE INCLUSIVE OF PLATING
3. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
4. MOLD FLASH SHALL NOT EXCEED 0.254mm
5. PACKAGE EIAJ REFERENCE IS SC-74A (EIAJ)
S6 Package
6-Lead Plastic SOT-23
(LTC DWG # 05-08-1634)
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